HFA3101 S E M I C O N D U C T O R Gilbert Cell UHF Transistor Array July 1995 Features Description * High Gain Bandwidth Product (fT) . . . . . . . . . . . 10GHz The HFA3101 is an all NPN transistor array configured as a Multiplier Cell. Based on Harris bonded wafer UHF-1 SOI process, this array achieves very high fT (10GHz) while maintaining excellent hFE and VBE matching characteristics that have been maximized through careful attention to circuit design and layout, making this product ideal for communication circuits. For use in mixer applications, the cell provides high gain and good cancellation of 2nd order distortion terms. * High Power Gain Bandwidth Product . . . . . . . . . 5GHz * Current Gain (hFE) . . . . . . . . . . . . . . . . . . . . Typically 70 * Low Noise Figure (Transistor) . . . . . . . . . . . . . . . 3.5dB * Excellent hFE and VBE Matching * Low Collector Leakage Current . . . . . . . . . . . . <0.01nA * Pin-to-Pin Compatible to UPA101 Ordering Information Applications PART NUMBER * Balanced Mixers PACKAGE * Multipliers HFA3101Y DIE * Demodulators/Modulators HFA3101B 8 Lead Plastic SOIC (N) * Automatic Gain Control Circuits HFA3101B96 8 Lead Plastic SOIC (N) - Tape and Reel * Phase Detectors * Fiber Optic Signal Processing * Wireless Communication Systems * Wide Band Amplification Stages * Radio and Satellite Communications * High Performance Instrumentation Pinout Q6 3 2 Q5 1 5 Q3 Q4 4 Q1 Q2 6 7 8 HFA3101 (SOIC) TOP VIEW NOTE: Q5 and Q6 - 2 Paralleled 3m x 50m Transistors Q1, Q2, Q3, Q4 - Single 3m x 50m Transistors CAUTION: These devices are sensitive to electrostatic discharge. Users should follow proper IC Handling Procedures. Copyright (c) Harris Corporation 1995 9-26 File Number 3663.2 Specifications HFA3101 Absolute Maximum Ratings Thermal Information VCEO, Collector to Emitter Voltage . . . . . . . . . . . . . . . . . . . . . . . 8.0V VCBO, Collector to Base Voltage . . . . . . . . . . . . . . . . . . . . . . . 12.0V VEBO, Emitter to Base Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . 5.5V IC, Collector Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30mA TSTG, Storage Temperature Range . . . . . . . . . . . . -65oC to +150oC Operating Temperature Range . . . . . . . . . . . . . . . . . -40oC to +85oC TJ, Junction Temperature (DIE) . . . . . . . . . . . . . . . . . . . . . . +175oC TJ, Junction Temperature (Plastic Package) . . . . . . . . . . . . +150oC Lead Temperature (Soldering 10s) (Lead Tips Only) . . . . . . +300oC Thermal Resistance JA Plastic 8 Lead SOIC Package . . . . . . . . . . . . . . . . . . . 185oC/W Maximum Package Power Dissipation at +75oC Plastic 8 Lead SOIC Package . . . . . . . . . . . . . . . . . . . . . . . . 0.4W Derating Factor Above +75oC Plastic 8 Lead SOIC Package . . . . . . . . . . . . . . . . . . . . 5.4mW/oC CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Electrical Specifications at +25oC PARAMETER TEST CONDITIONS (NOTE 1) TEST LEVEL ALL GRADES MIN TYP MAX UNITS Collector-to-Base Breakdown Voltage, V(BR)CBO, Q1 thru Q6 IC = 100A, IE = 0 A 12 18 - V Collector-to-Emitter Breakdown Voltage, V(BR)CEO, Q5 and Q6 IC = 100A, IB = 0 A 8 12 - V Emitter-to-Base Breakdown Voltage, V(BR)EBO, Q1 thru Q6 IE = 10A, IC = 0 A 5.5 6 - V Collector Cutoff Current, ICBO, Q1 thru Q4 VCB = 8V, IE = 0 A - 0.1 10 nA Emitter Cutoff Current, IEBO, Q5 and Q6 VEB = 1V, IC = 0 A - - 200 nA DC Current Gain, hFE, Q1 thru Q6 IC = 10mA, VCE = 3V A 40 70 - VCB = 5V, f = 1MHz C - 0.300 - pF - 0.600 - pF - 0.200 - pF - 0.400 - pF Collector-to-Base Capacitance, CCB Q1 thru Q4 Q5 and Q6 Emitter-to-Base Capacitance, CEB Q1 thru Q4 VEB = 0, f = 1MHz B Q5 and Q6 Q1 thru Q4 IC = 10mA, VCE = 5V C - 10 - GHz Q5 and Q6 IC = 20mA, VCE = 5V C - 10 - GHz Q1 thru Q4 IC = 10mA, VCE = 5V C - 5 - GHz Q5 and Q6 IC = 20mA, VCE = 5V C - 5 - GHz Available Gain at Minimum Noise Figure, GNFMIN, Q5 and Q6 IC = 5mA, VCE = 3 f = 0.5GHz C - 17.5 - dB f = 1.0GHz C - 11.9 - dB Minimum Noise Figure, NFMIN, Q5 and Q6 IC = 5mA, VCE = 3V f = 0.5GHz C - 1.7 - dB f = 1.0GHz C - 2.0 - dB IC = 5mA, VCE = 3V f = 0.5GHz C - 2.25 - dB f = 1.0GHz C - 2.5 - dB Current Gain-Bandwidth Product, fT Power Gain-Bandwidth Product, fMAX 50 Noise Figure, NF50, Q5 and Q6 DC Current Gain Matching, hFE1/hFE2, Q1 and Q2, Q3 and Q4, and Q5 and Q6 VCE = 3V, IC = 10mA A 0.9 1.0 1.1 Input Offset Voltage, VOS, (Q1 and Q2), (Q3 and Q4), (Q5 and Q6) IC = 10mA, VCE = 3V A - 1.5 5 mV Input Offset Current, IC, (Q1 and Q2), (Q3 and Q4), (Q5 and Q6) IC = 10mA, VCE = 3V A - 5 25 A Input Offset Voltage TC, dVOS/dT, (Q1 and Q2, Q3 and Q4, Q5 and Q6) IC = 10mA, VCE = 3V C - 0.5 - V/oC Collector-to-Collector Leakage, ITRENCH-LEAKAGE VTEST = 5V B - 0.01 - nA NOTE: 1. Test Level: A. Production Tested, B. Guaranteed Limit or Typical Based on Characterization, C. Design Typical for Information Only. 9-27 HFA3101 PSPICE Model for a 3m x 50m Transistor + (IS = 1.840E-16 XTI = 3.000E+00 EG = 1.110E+00 VAF = 7.200E+01 + VAR = 4.500E+00 BF = 1.036E+02 ISE = 1.686E-19 NE = 1.400E+00 + IKF = 5.400E-02 XTB = 0.000E+00 BR = 1.000E+01 ISC = 1.605E-14 + NC = 1.800E+00 IKR = 5.400E-02 RC = 1.140E+01 CJC = 3.980E-13 + MJC = 2.400E-01 VJC = 9.700E-01 FC = 5.000E-01 CJE = 2.400E-13 + MJE = 5.100E-01 VJE = 8.690E-01 TR = 4.000E-09 TF = 10.51E-12 + ITF = 3.500E-02 XTF = 2.300E+00 VTF = 3.500E+00 PTF = 0.000E+00 + XCJC = 9.000E-01 CJS = 1.689E-13 VJS = 9.982E-01 MJS = 0.000E+00 + RE = 1.848E+00 RB = 5.007E+01 RBM = 1.974E+00 KF = 0.000E+00 + AF = 1.000E+00) Common Emitter S-Parameters of 3m x 50m Transistor FREQ. (Hz) |S11| PHASE(S11) |S12| PHASE(S12) |S21| PHASE(S21) |S22| PHASE(S22) VCE = 5V and IC = 5mA 1.0E+08 0.83 -11.78 1.41E-02 78.88 11.07 168.57 0.97 -11.05 2.0E+08 0.79 -22.82 2.69E-02 68.63 10.51 157.89 0.93 -21.35 3.0E+08 0.73 -32.64 3.75E-02 59.58 9.75 148.44 0.86 -30.44 4.0E+08 0.67 -41.08 4.57E-02 51.90 8.91 140.36 0.79 -38.16 5.0E+08 0.61 -48.23 5.19E-02 45.50 8.10 133.56 0.73 -44.59 6.0E+08 0.55 -54.27 5.65E-02 40.21 7.35 127.88 0.67 -49.93 7.0E+08 0.50 -59.41 6.00E-02 35.82 6.69 123.10 0.62 -54.37 8.0E+08 0.46 -63.81 6.27E-02 32.15 6.11 119.04 0.57 -58.10 9.0E+08 0.42 -67.63 6.47E-02 29.07 5.61 115.57 0.53 -61.25 1.0E+09 0.39 -70.98 6.63E-02 26.45 5.17 112.55 0.50 -63.96 1.1E+09 0.36 -73.95 6.75E-02 24.19 4.79 109.91 0.47 -66.31 1.2E+09 0.34 -76.62 6.85E-02 22.24 4.45 107.57 0.45 -68.37 1.3E+09 0.32 -79.04 6.93E-02 20.53 4.15 105.47 0.43 -70.19 1.4E+09 0.30 -81.25 7.00E-02 19.02 3.89 103.57 0.41 -71.83 1.5E+09 0.28 -83.28 7.05E-02 17.69 3.66 101.84 0.40 -73.31 1.6E+09 0.27 -85.17 7.10E-02 16.49 3.45 100.26 0.39 -74.66 1.7E+09 0.25 -86.92 7.13E-02 15.41 3.27 98.79 0.38 -75.90 1.8E+09 0.24 -88.57 7.17E-02 14.43 3.10 97.43 0.37 -77.05 1.9E+09 0.23 -90.12 7.19E-02 13.54 2.94 96.15 0.36 -78.12 2.0E+09 0.22 -91.59 7.21E-02 12.73 2.80 94.95 0.35 -79.13 2.1E+09 0.21 -92.98 7.23E-02 11.98 2.68 93.81 0.35 -80.09 2.2E+09 0.20 -94.30 7.25E-02 11.29 2.56 92.73 0.34 -80.99 2.3E+09 0.20 -95.57 7.27E-02 10.64 2.45 91.70 0.34 -81.85 2.4E+09 0.19 -96.78 7.28E-02 10.05 2.35 90.72 0.33 -82.68 2.5E+09 0.18 -97.93 7.29E-02 9.49 2.26 89.78 0.33 -83.47 2.6E+09 0.18 -99.05 7.30E-02 8.96 2.18 88.87 0.33 -84.23 9-28 HFA3101 Common Emitter S-Parameters of 3m x 50m Transistor (Continued) FREQ. (Hz) |S11| PHASE(S11) |S12| PHASE(S12) |S21| PHASE(S21) |S22| PHASE(S22) 2.7E+09 0.17 -100.12 7.31E-02 8.47 2.10 88.00 0.33 -84.97 2.8E+09 0.17 -101.15 7.31E-02 8.01 2.02 87.15 0.33 -85.68 2.9E+09 0.16 -102.15 7.32E-02 7.57 1.96 86.33 0.33 -86.37 3.0E+09 0.16 -103.11 7.32E-02 7.16 1.89 85.54 0.33 -87.05 VCE = 5V and IC = 10mA 1.0E+08 0.72 -16.43 1.27E-02 75.41 15.12 165.22 0.95 -14.26 2.0E+08 0.67 -31.26 2.34E-02 62.89 13.90 152.04 0.88 -26.95 3.0E+08 0.60 -43.76 3.13E-02 52.58 12.39 141.18 0.79 -37.31 4.0E+08 0.53 -54.00 3.68E-02 44.50 10.92 132.57 0.70 -45.45 5.0E+08 0.47 -62.38 4.05E-02 38.23 9.62 125.78 0.63 -51.77 6.0E+08 0.42 -69.35 4.31E-02 33.34 8.53 120.37 0.57 -56.72 7.0E+08 0.37 -75.26 4.49E-02 29.47 7.62 116.00 0.51 -60.65 8.0E+08 0.34 -80.36 4.63E-02 26.37 6.86 112.39 0.47 -63.85 9.0E+08 0.31 -84.84 4.72E-02 23.84 6.22 109.36 0.44 -66.49 1.0E+09 0.29 -88.83 4.80E-02 21.75 5.69 106.77 0.41 -68.71 1.1E+09 0.27 -92.44 4.86E-02 20.00 5.23 104.51 0.39 -70.62 1.2E+09 0.25 -95.73 4.90E-02 18.52 4.83 102.53 0.37 -72.28 1.3E+09 0.24 -98.75 4.94E-02 17.25 4.49 100.75 0.35 -73.76 1.4E+09 0.22 -101.55 4.97E-02 16.15 4.19 99.16 0.34 -75.08 1.5E+09 0.21 -104.15 4.99E-02 15.19 3.93 97.70 0.33 -76.28 1.6E+09 0.20 -106.57 5.01E-02 14.34 3.70 96.36 0.32 -77.38 1.7E+09 0.20 -108.85 5.03E-02 13.60 3.49 95.12 0.31 -78.41 1.8E+09 0.19 -110.98 5.05E-02 12.94 3.30 93.96 0.31 -79.37 1.9E+09 0.18 -113.00 5.06E-02 12.34 3.13 92.87 0.30 -80.27 2.0E+09 0.18 -114.90 5.07E-02 11.81 2.98 91.85 0.30 -81.13 2.1E+09 0.17 -116.69 5.08E-02 11.33 2.84 90.87 0.30 -81.95 2.2E+09 0.17 -118.39 5.09E-02 10.89 2.72 89.94 0.29 -82.74 2.3E+09 0.16 -120.01 5.10E-02 10.50 2.60 89.06 0.29 -83.50 2.4E+09 0.16 -121.54 5.11E-02 10.13 2.49 88.21 0.29 -84.24 2.5E+09 0.16 -122.99 5.12E-02 9.80 2.39 87.39 0.29 -84.95 2.6E+09 0.15 -124.37 5.12E-02 9.49 2.30 86.60 0.29 -85.64 2.7E+09 0.15 -125.69 5.13E-02 9.21 2.22 85.83 0.29 -86.32 2.8E+09 0.15 -126.94 5.13E-02 8.95 2.14 85.09 0.29 -86.98 2.9E+09 0.15 -128.14 5.14E-02 8.71 2.06 84.36 0.29 -87.62 3.0E+09 0.14 -129.27 5.15E-02 8.49 1.99 83.66 0.29 -88.25 9-29 HFA3101 Typical Performance Curves for Transistors 70 x 10-3 Ib = 200 Ib = 800 Ib = 400 Ib. = 1m Ib = 600 VCE = 5 140 120 60 100 50 hFE IC (A) 80 40 60 30 40 20 20 10 0 0 10-10 0 2.0 4.0 10-8 10-6 6.0 VCE (V) FIGURE 1. IC vs VCE VCE = 3 12 10-2 x 109 8 fT (Hz) IC & IB (A) 100 10 10-4 10-6 6 10-8 4 10-10 2 10-12 0.20 10-2 FIGURE 2. HFE vs IC 0.40 0.60 VBE (V) 0.80 0 10-4 1.0 10-2 IC (A) FIGURE 3. GUMMEL PLOT NOISE FIGURE (dB) 10-3 FIGURE 4. FT vs IC 4.8 20 4.6 18 4.4 16 4.2 14 4.0 12 3.8 |S21| (dB) 100 10-4 IC (A) 10 3.6 6 3.4 3.2 0 0.5 1.0 1.5 2.0 2.5 4 3.0 FREQUENCY (GHz) FIGURE 5. GAIN AND NOISE FIGURE vs FREQUENCY NOTE: Figures 14 through 18 are only for Q5 and Q6. 9-30 10-1 HFA3101 Die Characteristics PROCESS UHF-1 DIE DIMENSIONS: 53 x 52 x 14 1mils 1340m x 1320m x 355.6m 25.4m METALLIZATION: Type: Metal 1: AlCu(2%)/TiW Thickness: Metal 1: 8kA 0.5kA Type: Metal 2: AlCu(2%) Thickness: Metal 2: 16kA 0.8kA GLASSIVATION: Type: Nitride Thickness: 4kA 0.5kA DIE ATTACH: Material: Epoxy WORST CASE CURRENT DENSITY: 1.3636 x 105 A/cm2 Metallization Mask Layout HFA3101 7 7 6 6 8 5 8 5 1 4 1 4 2 2 3 9-31 3 HFA3101 Application Information The HFA3101 array is a very versatile RF Building block. It has been carefully laid out to improve its matching properties, bringing the distortion due to area mismatches, thermal distribution, betas and ohmic resistances to a minimum. The cell is equivalent to two differential stages built as two "variable transconductance multipliers" in parallel, with their outputs cross coupled. This configuration is well known in the industry as a Gilbert Cell which enables a four quadrant multiplication operation. Due to the input dynamic range restrictions for the input levels at the upper quad transistors and lower tail transistors, the HFA3101 cell has restricted use as a linear four quadrant multiplier. However, its configuration is well suited for uses where its linear response is limited to one of the inputs only, as in modulators or mixer circuit applications. Examples of these circuits are up converters, down converters, frequency doublers and frequency/phase detectors. Figure 6 shows the typical input waveforms where the frequency of the carrier is higher than the modulating signal. The output waveform shows a typical suppressed carrier output of an up converter or an AM signal generator. Carrier suppression capability is a property of the well known Balanced modulator in which the output must be zero when one or the other input (carrier or modulating signal) is equal to zero. however, at very high frequencies, high frequency mismatches and AC offsets are always present and the suppression capability is often degraded causing carrier and modulating feedthrough to be present. Being a frequency translation circuit, the balanced modulator has the properties of translating the modulating frequency (M) to the carrier frequency (C), generating the two side bands U = C + M and L = C - M. Figure 7 shows some translating schemes being used by balanced mixers. C - M Although linearization is still an issue for the lower pair input, emitter degeneration can be used to improve the dynamic range and consequent linearity. The HFA3101 has the lower pair emitters brought to external pins for this purpose. In modulators applications, the upper quad transistors are used in a switching mode where the pairs Q1/Q2 and Q3/Q4 act as non saturating high speed switches. These switches are controlled by the signal often referred as the carrier input. The signal driving the lower pair Q5/Q6 is commonly used as the modulating input. This signal can be linearly transferred to the output by either the use of low signal levels (Well below the thermal voltage of 26mV) or by the use of emitter degeneration. The chopped waveform appearing at the output of the upper pair (Q1 to Q4) resembles a signal that is multiplied by +1 or -1 at every half cycle of the switching waveform. C + M C FIGURE 7A. UP CONVERSION OR SUPPRESSED CARRIER AM IF (C - M) FOLDED BACK M CARRIER SIGNAL +1 C -1 FIGURE 7B. DOWN CONVERSION MODULATING SIGNAL C BASEBAND M DIFFERENTIAL OUTPUT FIGURE 7C. ZERO IF OR DIRECT DOWN CONVERSION FIGURE 7. MODULATOR FREQUENCY SPECTRUM FIGURE 6. TYPICAL MODULATOR SIGNALS 9-32 HFA3101 The use of the HFA3101 as modulators has several advantages when compared to its counterpart, the diode doublebalanced mixer, in which it is required to receive enough energy to drive the diodes into a switching mode and has also some requirements depending on the frequency range desired, of different transformers to suit specific frequency responses. The HFA3101 requires very low driving capabilities for its carrier input and its frequency response is limited by the FT of the devices, the design and the layout techniques being utilized. Up conversion uses, for UHF transmitters for example, can be performed by injecting a modulating input in the range of 45MHz to 130MHz that carries the information often called IF (Intermediate frequency) for up conversion (The IF signal has been previously modulated by some modulation scheme from a baseband signal of audio or digital information) and by injecting the signal of a local oscillator of a much higher frequency range from 600MHz to 1.2GHz into the carrier input. Using the example of a 850MHz carrier input and a 70MHz IF, the output spectrum will contain a upper side band of 920MHz, a lower side band of 780MHz and some of the carrier (850MHz) and IF (70MHz) feedthrough. A Band pass filter at the output can attenuate the undesirable signals and the 920MHz signal can be routed to a transmitter RF power amplifier. Down conversion, as the name implies, is the process used to translate a higher frequency signal to a lower frequency range conserving the modulation information contained in the higher frequency signal. One very common typical down conversion use for example, is for superheterodyne radio receivers where a translated lower frequency often referred as intermediate frequency (IF) is used for detection or demodulation of the baseband signal. Other application uses include down conversion for special filtering using frequency translation methods. An oscillator referred as the local oscillator (LO) drives the upper quad transistors of the cell with a frequency called C . The lower pair is driven by the RF signal of frequency M to be translated to a lower frequency IF. The spectrum of the IF output will contain the sum and difference of the frequencies C and M. Notice that the difference can become negative when the frequency of the local oscillator is lower than the incoming frequency and the signal is folded back as in Figure 7. NOTE: The acronyms RF, IF and LO are often interchanged in the industry depending on the application of the cell as mixers or modulators. The output of the cell also contains multiples of the frequency of the signal being fed to the upper quad pair of transistors because of the switching action equivalent to a square wave multiplication. In practice, however, not only the odd multiples in the case of a symmetrical square wave but some of the even multiples will also appear at the output spectrum due to the nature of the actual switching waveform and high frequency performance. By-products of the form M*C + N*M with M and N being positive or negative integers are also expected to be present at the output and their levels are carefully examined and minimized by the design. This distortion is considered one of the figures of merit for a mixer application. The process of frequency doubling is also understood by having the same signal being fed to both modulating and carrier ports. The output frequency will be the sum of C and M which is equivalent to the product of the input frequency by 2 and a zero Hz or DC frequency equivalent to the difference of C and M . Figure 7 also shows one technique in use today where a process of down conversion named zero IF is made by using a local oscillator with a very pure signal frequency equal to the incoming RF frequency signal that contains a baseband (audio or digital signal) modulation. Although complex, the extraction or detection of the signal is straightforward. Another useful application of the HFA3101 is its use as a high frequency phase detector where the two signals are fed to the carrier and modulation ports and the DC information is extracted from its output. In this case, both ports are utilized in a switching mode or overdrive, such that the process of multiplication takes place in a quasi digital form (2 square waves). One application of a phase detector is frequency or phase demodulation where the FM signal is split before the modulating and carrier ports. The lower input port is always 90 degrees apart from the carrier input signal through a high Q tuned phase shift network. The network, being tuned for a precise 90 degrees shift at a nominal frequency, will set the two signals 90 degrees apart and a quiescent output DC level will be present at the output. When the input signal is frequency modulated, the phase shift of the signal coming from the network will deviate from 90 degrees proportional to the frequency deviation of the FM signal and a DC variation at the output will take place, resembling the demodulated FM signal. The HFA3101 could also be used for quadrature detection, (I/Q demodulation), AGC control with limited range, low level multiplication to name a few other applications. Biasing Various biasing schemes can be employed for use with the HFA3101. Figure 8 shows the most common schemes. The biasing method is a choice of the designer when cost, thermal dependence, voltage overheads and DC balancing properties are taken into consideration. Figure 8A shows the simplest form of biasing the HFA3101. The current source required for the lower pair is set by the voltage across the resistor RBIAS less a VBE drop of the lower transistor. To increase the overhead, collector resistors are substituted by a RF choke as the upper pair functions as a current source for AC signals. The bases of the upper and lower transistors are biased by RB1 and RB2 respectively. The voltage drop across the resistor R2 must be higher than a VBE with an increase sufficient to assure that the collector to base junctions of the lower pair are always reverse biased. Notice that this same voltage also sets the VCE of operation of the lower pair which is important for the optimization of gain. Resistors REE are nominally zero for applications where the input signals are well below 25mV peak. Resistors REE are used to increase the linearity of the circuit upon higher level signals. The drop across REE must be taken into consideration when setting the current source value. Figure 8B depicts the use of a common resistor sharing the current through the cell which is used for temperature compensation as the lower pair VBE drop at the rate of -2mV/oC. Figure 8C uses a split supply. 9-33 HFA3101 VCC VCC RC VCC LCH LCH Q1 Q2 5 6 7 5 8 RB1 R2 Q3 Q4 Q5 LCH R1 6 8 5 6 7 R1 RB1 7 RB1 8 R1 Q1 Q2 Q6 Q5 R2 Q1 Q2 Q3 Q4 Q6 Q3 Q4 Q6 Q5 REE REE REE RBIAS REE REE 4 3 2 1 4 3 2 1 4 3 2 1 R2 REE RBIAS RBIAS RB2 RB2 RB2 RE RE RE VEE FIGURE 8A. FIGURE 8B. FIGURE 8C. FIGURE 8. Design Example: Down Converter Mixer Figure 9 shows an example of a low cost mixer for cellular applications. VCC 3V LO IN 0.1 LCH 390nH 0.01 2K IF OUT The choice of the bias has been related to the available power supply voltage with the values of R1, R2 and RBIAS splitting the voltages for optimum VCE values. For evaluation of the cell quiescent currents, the voltage at the emitter resistor RE has been recorded. 5p TO 12p 5 6 7 825MHz VCC 8 51 75MHz 0.01 110 Q1 Q2 Q3 Q4 Q5 The gain of the circuit, being a function of the load and the combined emitter resistances at high frequencies have been kept to a maximum by the use of an output match network. The high output impedance of the HFA3101 permits broadband match if so desired at 50 (RL = 50 to 2k) as well as with tuned medium Q matching networks (L, T etc.). Q6 4 3 2 330 1 RF IN 0.01 51 0.01 0.01 900MHz 220 the evaluation of the same layout for various quiescent current values and lower supply voltages. The choice of RE became important for the available overhead and also for maintaining an AC true impedance for high frequency signals. The value of 27 has been found to be the optimum minimum for the application. The input impedances of the HFA3101 base input ports are high enough to permit their termination with 50 resistors. Notice the AC termination by decoupling the bias circuit through good quality capacitors. Stability 27 VEE FIGURE 9. 3V DOWN CONVERTER APPLICATION The design flexibility of the HFA3101 is demonstrated by a low cost, and low voltage mixer application at the 900MHz range. The choice of good quality chip components with their self resonance outside the boundaries of the application are important. The design has been optimized to accommodate The cell, by its nature, has very high gain and precautions must be taken to account for the combination of signal reflections, gain, layout and package parasitics. The rule of thumb of avoiding reflected waves must be observed. It is important to assure good matching between the mixer stage and its front end. Laboratory measurements have shown some susceptibility for oscillation at the upper quad transistors input. Any LO prefiltering has to be designed such the return loss is maintained within acceptable limits specially at high frequencies. Typical off the shelf filters exhibits very 9-34 HFA3101 poor return loss for signals outside the passband. It is suggested that a "pad" or a broadband resistive network be used to interface the LO port with a filter. The inclusion of a parallel 2K resistor in the load decreases the gain slightly which improves the stability factor and also improves the distortion products (output intermodulation or 3rd order intercept). The employment of good RF techniques shall suffice the stability requirements. TABLE 2. S22 PARAMETERS FOR DOWN CONVERSION, LCH = 10H LO LEVEL VCC = 3V IBIAS = 8mA Power Gain -6dBm 8.5dB TOI Output -6dBm 11.5dBm NF SSB -6dBm 14.5dB Evaluation Power Gain 0dBm 8.6dB The evaluation of the HFA3101 in a mixer configuration is presented in Figures 11 to Figure 16, Table 1 and Table 2. The layout is depicted in Figure 10. TOI Output 0dBm 11dBm NF SSB 0dBm 15dB LO LEVEL VCC = 4V IBIAS = 19mA Power Gain -6dBm 10dB TOI Output -6dBm 13dBm NF SSB -6dBm 20dB Power Gain 0dBm 11dB TOI Output 0dBm 12.5dBm NF SSB 0dBm 24dB PARAMETER PARAMETER TABLE 3. TYPICAL VALUES OF S22 FOR THE OUTPUT PORT. LCH = 390nH IBIAS = 8mA (SET UP OF FIGURE 11) FREQUENCY RESISTANCE REACTANCE 300MHz 22 -115 600MHz 7.5 -43 900MHz 5.2 -14 1.1GHz 3.9 0 TABLE 4. TYPICAL VALUES OF S22. LCH = 390nH, IBIAS = 18mA FIGURE 10. UP/DOWN CONVERTER LAYOUT, 400%. MATERIAL G10, 0.031 The output matching network has been designed from data taken at the output port at various test frequencies with the setup as in Table 1. S22 characterization is enough to assure the calculation of L, T or transmission line matching networks. TABLE 1. S22 PARAMETERS FOR DOWN CONVERSION, LCH = 10H FREQUENCY RESISTANCE REACTANCE 10MHz 265 615 45MHz 420 - 735 75MHz 122 - 432 100MHz 67 - 320 FREQUENCY RESISTANCE REACTANCE 300MHz 23.5 -110 600MHz 10.3 -39 900MHz 8.7 -14 1.1GHz 8 0 Up Converter Example An application for a up converter as well as a frequency multiplier can be demonstrated using the same layout, with an addition of matching components. The output port S22 must be characterized for proper matching procedures and depending on the frequency desired for the output, transmission line transformations can be designed. The return loss of the input ports maintain acceptable values in excess of 1.2GHz which can permit the evaluation of a frequency doubler to 2.4GHz if so desired. The addition of the resistors REE can increase considerably the dynamic range of the up converter as demonstrated at Figure 18. The evaluation results depicted in Table 5 have been obtained by a triple stub tuner as a matching network for the output due to the layout constraints. Based on the evaluation results it is clear that the cell requires a higher Bias current for overall performance. 9-35 HFA3101 VCC 3V LCH S11 LOG MAG 0dB 5dB/DIV 0.1 2K 5 6 7 8 4V 3V Q6 4 2 1 Q5 Q3 Q4 3 Q1 Q2 100MHz FIGURE 11. OUTPUT PORT S22 TEST SET UP FIGURE 12. LO PORT RETURN LOSS S22 LOG MAG S11 LOG MAG 0dB 10dB/DIV 0dB 5dB/DIV 100MHz 10MHz 1.1GHz FIGURE 13. RF PORT RETURN LOSS 110MHz FIGURE 14. IF PORT RETURN LOSS, WITH MATCHING NETWORK RF = 900MHz -25dBm LO = 825MHz -6dBm RF = 901MHz - 25dBm LO = 825MHz -6dBm 10dB/ DIV 1.1GHz 10dB/ DIV -17dBm -26dBm -36dBm -53dBm -58dBm 64M 11*LO - 10RF 76MHz IF 88M 12RF - 13LO SPAN 500MHz SPAN 40MHz 675 750 LO - 2RF FIGURE 15. TYPICAL IN BAND OUTPUT SPECTRUM, VCC = 3V 9-36 825 900 975 LO + 2RF FIGURE 16. TYPICAL OUT OF BAND OUTPUT SPECTRUM HFA3101 TABLE 5. TYPICAL PARAMETERS FOR AN UP CONVERTER EXAMPLE Design Example: Up Converter Mixer Figure 17 shows an example of a up converter for cellular applications. VCC = 3V IBIAS = 8mA VCC = 4V IBIAS = 18mA Power Gain, LO = -6dBm 3dB 5.5dBm Power Gain, LO = 0dBm 4dB 7.2dB RF Isolation, LO = 0dBm 15dBc 22dBc LO Isolation, LO = 0dBm 28dBc 28dBc PARAMETER Conclusion The HFA3101 offers the designer a number of choices and different applications as a powerful RF building block. Although isolation is degraded from the theoretical results for the cell due to the unbalanced, nondifferential input schemes being used, a number of advantages can be taken into consideration like cost, flexibility, low power and small outline when deciding for a design. VCC 3V 0.1 LO IN 0.01 390nH 825MHz 0.01 5.2nH 900MHz 51 5 6 3V 7 VCC 8 11p 0.01 110 Q1 Q2 Q5 Q3 Q4 Q6 0.01 REE 4 3 2 1 0.01 330 REE RF IN 75MHz 51 0.01 220 27 FIGURE 17. OUTPUT WITHOUT EMITTER DEGENERATION 890 2LO - 10RF RF = 76MHz LO = 825MHz 901 912 12RF OUTPUT WITH EMITTER DEGENERATION REE = 4.7 EXPANDED SPECTRUM REE = 4.7 825 SPAN 50MHz FIGURE 18. TYPICAL SPECTRUM PERFORMANCE 9-37 900 976